Digital multi-band predistortion linearizer with non-linear subsampling algorithm in the feedback loop

ABSTRACT

A concurrent multi-band linearized transmitter (CMLT) has a concurrent d a multi-band predistortion block (CDMPB) and a concurrent multi-band transmitter (CMT) connected to the CDMPB, The CDMPB can have a plurality of digital baseband signal predistorter blocks (DBSPBs), an analyzing and modeling (A&amp;M) stage, and a signal observation feedback loop. Each DBSPB can have a plurality of inputs, each corresponding to a single frequency band of the multi-band input signal, and its output corresponding to a single frequency band; each output connect corresponding to an input of the CMLT. The A&amp;M stage can have a plurality of outputs connected to and updating the parameters of the DBSPBs, and a plurality of inputs connected to either both outputs of the signal observation loop or the output of the subsampling loop and to outputs of the DBSPBs. The A&amp;M stage can perform signals&#39; time alignment, reconstruction of signals and compute parameters of DBSPBs.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a Continuation of U.S. Ser. No. 14/467,642 filedAug. 25, 2014 which is a Continuation of U.S. Ser. No. 13/274,290 filedOct. 14, 2011, all of which are in their entirety incorporated herein byreference.

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH

Not Applicable.

APPENDIX

Not Applicable.

BACKGROUND OF THE INVENTION Field of the Invention

The present invention relates to multi-band digital predistortionlinearization.

Related Art

Power-efficient, low-complex, and reconfigurable radio system requiresthe design of energy-efficient transmitter and receiver architectures.At the transmitter side, the power consumption is mainly dominated bythe RF power amplification (PA) unit. Generally, PAs are the most powerconsuming and the least power efficient components of the RF chain.Moreover, their nonlinear behavior and non-flat frequency responseintroduce unwanted intermodulation distortions into the system, whichcould significantly degrade the output signal quality.

An efficient and proper approach to linearize the transmitternonlinearities, including the frequency up-conversion and poweramplification units, is digital predistortion (DPD) linearizationtechnique. The DPD technique is based on developing a reverse model ofthe nonlinear behavior and predistorted the input signals accordingly inorder to compensate for the distortions and nonlinearities introduced bythe transmitter.

In dual-band system, the nonlinear behavior of the device will introduceintermodulation, cross modulation, and harmonic products caused by thetwo fundamental signals. This can be extended to multi-band systemswhere more than two active signals are transmitted simultaneously.

The linearization of multi-band transmitter is based on the digitalpredistortion linearization. The DPD technique compensates for thetransmitter nonlinearity while operating in the high efficiency andnonlinear region. As an example presented here in this patent, twosignal processing blocks are employed to deal and compensate for theunwanted distortions and intermodulation products of the dual-bandtransmitter. In the scenario of multi-band transmitter (dual-band ormore) this processing architecture can be expanded to multipleprocessing block for linearization and distortion compensation ofmulti-band transmitter.

In order to obtain samples of the signal from the output of themulti-band system, multi-branch or multi-band down converter is requiredin the feedback loop. This feedback loop can be developed usingmulti-band down conversion unit, multi-branch down conversion unit, orusing subsampling based down conversion unit.

In one case, an energy-efficient and low-complex subsampling receiver isadopted in the feedback loop of the multi-band linearizationarchitecture. The subsampling receiver architecture is designed toconcurrently down-convert the multiple RF signals through singlereceiver chain. Using subsampling technique simplifies the feedback looptopology, requires fewer number of RF components, and reduces the powerconsumption.

Substituting the multi-band or multi-branch receiver feedback loop ofthe linearization topology with subsampling receiver architecturereduces the complexity of the system. The subsampling down conversion isnot very common as receivers because of its insufficient performance inthe presence of uncontrolled interfering signals. However, in the caseof a DPD feedback loop, the problem is different and the interferingsignals can be controlled such that they will not affect the signalquality. The different intermodulation, cross modulation and harmonicproducts make choosing the sampling frequency a complex task in order toavoid any overlap between the down-converted desired signals and theirintermodulation and cross modulation products. Therefore, it isimperative to develop an algorithm to select the sampling frequency sothat it takes into account all the possible frequencies such that thetarget signals will not be interfered with the undesired product terms.

Summary of the Invention

In one aspect of the present invention, a concurrent digital multi-bandlinearizer compromises a baseband signal preprocessing block, thebaseband signal processing block including a digital predistortion unit;a signal up-conversion block, an RF power amplification block, the RFpower amplification block including the concurrent multi-band poweramplifier; and an RF power combining network.

In the description of the invention, a concurrent dual-band amplifierwill be used. It is noted that a concurrent dual-band power amplifierwhich includes one amplification unit for two frequency bands may beconsidered in one sense a simple and special case of multi-band poweramplifiers.

In one aspect of the present invention, the feedback loop of the digitalpredistortion consists of multiple RF down-conversion units associatedwith each of the frequency band of operation.

In another aspect of the present invention, a single feedback loop basedon subsampling receiver technique is used to down-convert and extractthe RF signals form all the frequency band at the same time.

Further areas of applicability of the present invention will becomeapparent with reference to the following drawings, description andclaims. It should be understood that the detailed description andspecific examples, while indicating the preferred embodiment of theinvention, are intended for purposes of illustration only and are notintended to limit the scope of the invention.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention will become more fully understood from thedetailed description and the accompanying drawings, wherein:

FIG. 1 is a block diagram of the dual-band digital predistortionarchitecture according to an exemplary embodiment of the presentinvention.

FIG. 2 is an alternate embodiment illustrating a detailed block diagramof the architecture of FIG. 1, using subsampling based feedback loop.

FIG. 3 is one embodiment illustrating a detailed block diagram of thesubsampling receiver architecture of FIG. 2.

FIG. 4A is the fundamental signal representation at the input of thedual-band transmitter.

FIG. 4B is the fundamental signal representation at the output of thedual-band transmitter and intermodulation terms at the output of thenonlinear transmitter.

FIG. 5 is dual-band RF signal and the frequency position of thesubsampling harmonics.

FIG. 6 is a flowchart illustrating the steps of the execution of findingthe possible subsampling frequencies.

FIG. 7A is the power spectrum of the predistortion results for 880 MHz,using dual-band digital prediction technique with dual-branch feedbackloop. Spectrums marked with circles are the output withoutlinearization, those marked with diamonds are the output afterlinearization, and those marked with solid rectangles are the inputssignals.

FIG. 7B is the power spectrum of the predistortion results for 1978 MHz,using dual-band digital prediction technique with dual-branch feedbackloop, using the same codes as described for FIG. 7A.

FIG. 8 is the Power spectrum of the output of the dual-band nonlineartransmitter including the two fundamental RF signals and theirinter-modulation products and harmonics.

FIG. 9 is the Power spectrum of the captured signal using an ADCoperating at 619.8 MHz sampling frequency.

FIG. 10A is the power spectrum of the predistortion results for 880 MHz.using dual-band digital prediction technique with subsampling feedbackloop.

FIG. 10B is the power spectrum of the predistortion results for 1978MHz, using dual-band digital prediction technique with subsamplingfeedback loop.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The following description of the preferred embodiment(s) is merelyexemplary in nature and is in no way intended to limit the invention,its application, or uses.

Broadly, an embodiment of the present invention provides multiple branchdigital predistortion linearization architecture and digital signalprocessing algorithms for impairments-free operation and linearizedmulti-band transmitter.

Referring to FIG. 1, the system block diagram of the dual-bandlinearization architecture 100 is displayed. The input signals, x1 andx2, 105 are fed into two distinct predistorters blocks 110. Thepredistorted signals 115 are converted from digital to analog 120 andup-converted 125 to RF frequencies. Then the two RF signals are combined130 and amplified by the power amplifier 135.

For digital predistortion linearization and identify the inverse model,the sample of the RF signal are captured using dual-band coupler 140.Then the RF signals are bandpass filtered 145, frequency down converted150, digitized using analog-to-digital converters 155. The digitaloutput samples 160, the input signals 105 and predistorted signals 115are used in the analyzing stage 165 for nonlinear model identificationand reverse modeling.

The feedback path of the dual-band linearizer requires the use of twodown-conversion stages 150, as well as bandpass fitters 145 to removemost of the imperfections caused by the power amplifier. Thepredistorted inputs, x_(pd1) and x X_(pd2), 115 as well as the output ofeach band of the PA, y₁ and y₂, 160 are used to generate thepredistorter signal processing model 110. The processing model equationsof the linearization processing algorithm 165 for prediction andcompensation of the distortions and intermodulations is as follows:

$\begin{matrix}{{{x_{{pd}\; 1}(n)} = {\sum\limits_{m = 0}^{M - 1}\; {\sum\limits_{k = 0}^{K - 1}\; {\sum\limits_{j = 0}^{k}\; {c_{1,j,k,m}{x_{1}\left( {n - m} \right)}{{x_{1}\left( {n - m} \right)}}^{k - j}{{x_{2}\left( {n - m} \right)}}^{j}}}}}}{{x_{{pd}\; 2}(n)} = {\sum\limits_{m = 0}^{M - 1}\; {\sum\limits_{k = 0}^{K - 1}\; {\sum\limits_{j = 0}^{k}\; {c_{2,j,k,m}{x_{2}\left( {n - m} \right)}{{x_{2}\left( {n - m} \right)}}^{k - j}{{x_{1}\left( {n - m} \right)}}^{j}}}}}}} & (1)\end{matrix}$

Where x₁(n) and x₂(n) are the input signals, x_(pd1)(n) and x_(pd2)(n)are the predistorted signals to the input of the dual-band transmitter,c_(1,j,k,m) and c_(2,j,k,m) are the identified model's coefficients, andfinally M is the order of the memory effect and K is the order ofnonlinearity.

Concurrent multi-band receiver architectures require a bandpass filter145, down-conversion stage 150, and ADC 155 for the translation of eachRF frequency bands to baseband. Using subsampling with a high speed ADCallows the elimination of all these components; however, the user needsto make sure that the signals don't overlap in the subsampled spectraldomain.

Sampling multi-bands at the same time also eliminates the time delaytaken between different band paths caused by the filters. FIG. 2displays the dual-band predistortion architecture with a subsamplingfeedback loop 200. At the feedback loop, it consists of optionalbandpass filters 245, a track and hold 250, an analog-to-digitalconverter 255, a digital conversion unit 260, and analyzing stage 270.

Sampling the band-limited RF signal at frequency rates much lower thanthe carrier frequency, but higher than signal bandwidth folds the RFsignal to the lower frequencies, where these replicates of the RF signalat baseband or intermediate frequencies can he used to reconstruct thebaseband signal. To make sure that there is no aliasing between thereplicas, the subsampling rate should be chosen in the following range:

$\begin{matrix}{{\frac{2\; f_{U}}{n} \leq f_{s} \leq {\frac{2\; f_{L}}{n - 1}\mspace{14mu} {where}\mspace{14mu} 1} \leq n \leq \left\lfloor \frac{f_{U}}{B} \right\rfloor}{f_{s} \geq {2 \times B\mspace{14mu} {Nyquist}\mspace{14mu} {rate}}}} & (2)\end{matrix}$

where f_(L) and f_(U) are the lower and upper frequencies of theband-limited RF signal, B=f_(U)−f_(L) is the signal bandwidth, and n isan integer value.

FIG. 3 shows a general block diagram of subsampling-based receiver 300.It consists of RF bandpass filter 305, low-noise amplifier (LNA) 310,subsampling receiver 345 including the track and hold (T&H) 320, and ADC325 followed by baseband digital signal processing (DSP) unit 330. TheT&H 320 is required to expand the analog bandwidth of the receiver anddefines the RF range of receiver operation. The sampling clock of theT&H 320 and ADC 325 are chosen from (2) to avoid any aliasing with theother RF signals.

In dual-band operation transmitter with nonlinearity, the first aridsecond bands will produce intermodulation, cross modulation and harmonicproducts. FIG. 4 shows the power spectrum of the input signal, and theoutput signal when passed through a third order nonlinear system. Theinput signals as two-tone signal around carrier frequencies of ω₁ and ω₂(FIG. 4(a)) the output signals are around carrier frequencies of ω₁ andω₂ and the 3rd order intermodulation frequencies of 2ω₁−ω₁ and 2ω₂−ω₁(FIG. 4(b)). The unwanted intermodulation can he classified into threegroups of cross-modulation, in-band intermodulation, and out-of-bandintermodulation terms. This latter is usually filtered out beforetransmission using RF filter to avoid signal quality degradation andinterference with the signals at the adjacent channels.

Now considering two RF signals at carrier frequencies of ω₂ and ω₂, withtheir respective bandwidths B1 and B2 as shown in FIG. 5, thesubsampling frequency, fs, must be chosen to ensure that the two signalsdo not overlap in the subsampled domain. Taking into account thesubsampling theorem for sampling the multi-band signals and followingthe neighbor and boundary constraints, an iterative process is used tofind all the valid subsampling frequencies for the two fundamentalfrequencies.

The, out-of-band intermodulation-modulation, and harmonics generated bythe fundamental signals are not required for the predistortionapplication; therefore, an iterative subsampling algorithm has beendeveloped to subsample the RF signals without any overlap with the otherunwanted RF signals. FIG. 6 is the flowchart of the developed iterativesubsampling algorithm to find the valid subsampling frequencies so thatthe replicas of the wanted RF signals have no overlap with the harmonicsand intermodulation frequency terms.

Referring to FIG. 7, there is shown the measured output spectrum of thedual-band transmitter 170 for three cases: 1) without using thedual-band digital pre-compensator 110, 2) using the dual-band digitalpre-compensator 110 3) the input signal. The output spectrum of case-2with digital pre-compensator shows that the digital pre-compensator 110can compensate for the cross-modulation and in-band inter-modulationterms introduced by the transmitter nonlinearity.

Referring to FIG. 8, there is an example of the power spectrum at theoutput of the concurrent dual-band nonlinear transmitter which containstwo fundamental RF frequencies and their corresponding harmonics andinter-modulation products.

As an example for the application of this invention, FIG. 9, points upthe power spectrum after the subsampling feedback loop which shows thetwo fundamental RF signals at 260.2 MHz and 118.6 MHz when thesubsampling frequency of 619.8 is used following is determination by thedeveloped iterative subsampling algorithm illustrated in FIG. 6. Thespectrum in FIG. 9 shows that the harmonics and inter-modulation termshave no interference with the two desired fundamental signals.

Referring to FIG. 10, there is shown the measured output spectrum of thedual-band transmitter 280 for three cases: 1) without using thedual-band digital pre-compensator 210, 2) using the dual-band digitalpre-compensator 210, 3) the input signal. The output spectrum of case-2with digital pre-compensator shows that the digital pre-compensator 210can compensate for the cross-modulation and in-band inter-modulationterms introduced by the transmitter dynamic nonlinearity with memoryeffects.

What is claimed is:
 1. A concurrent multi-band linearized transmittercomprising: a concurrent digital multi-band predistortion block; aconcurrent multi-band linearized transmitter connected to saidconcurrent digital multiband predistortion block.
 2. The concurrentmulti-band linearized transmitter of claim 1, wherein said concurrentdigital multi-band predistortion block further comprises: a plurality ofdigital baseband signal predistorter blocks; an analyzing and modellingstage; and a signal observation feedback loop,
 3. The concurrentmulti-band linearized transmitter of claim 2, wherein said plurality ofdigital baseband signal predistorter blocks further comprises: aplurality of inputs, each input corresponding to a single frequencychannel; a plurality of outputs, each output corresponding to a singlefrequency channel, and each output connected to an input of saidconcurrent multi-band transmitter.
 4. The concurrent multi-bandlinearized transmitter of claim 2, wherein said analyzing and modelingstage further comprises: a plurality of outputs connected to andupdating the parameters of said digital baseband signal predistorterblock; a plurality of inputs connected to both (a) said outputs of saidsignal observation feedback loop, and (b) to said outputs of saiddigital baseband signal predistorter blocks.
 5. The concurrentmulti-band linearized transmitter of claim 2, wherein said analyzing andmodeling stage is further adapted to: perform time alignment of complexbaseband signals from sampling said outputs of said concurrentmulti-band transmitter; and perform the reconstruction of the complexbaseband signals from sampling said outputs of said concurrentmulti-band transmitter.
 6. The concurrent multi-band linearizedtransmitter of claim 2, wherein said signal observation feedback loopfurther is further adapted to: down-convert samples of the RF signals atsaid output of the concurrent multi-band transmitter of claim 1; andextract from said down-converted samples a baseband equivalent for allfrequency channels.
 7. The concurrent multi-band linearized transmitterof claim 2, wherein said signal observation feedback loop furthercomprises for each channel an RF filter; A signal down conversion block;and an analog-to-digital converter (ADC).
 8. The concurrent multi-bandlinearized transmitter of claim 2, wherein said signal observationfeedback loop further comprises: a single subsampling-based receiver todown-convert samples output from a concurrent multi-band transmitter. 9.The concurrent multi-band linearized transmitter of claim 8, whereinsaid single subsampling -based receiver further comprises: an RF filter;a track and hold (T&H) block; and an analog-to-digital converter (ADC).